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FEATURES Enhanced Replacement for LF347 and TL084 AC PERFORMANCE 1 s Settling to 0.01% for 10 V Step 20 V/ s Slew Rate 0.0003% Total Harmonic Distortion (THD) 4 MHz Unity Gain Bandwidth DC PERFORMANCE 0.5 mV max Offset Voltage (AD713K) 20 V/C max Drift (AD713K) 200 V/mV min Open Loop Gain (AD713K) 2 V p-p typ Noise, 0.1 Hz to 10 Hz True 14-Bit Accuracy Single Version: AD711, Dual Version: AD712 Available in 16-Pin SOIC, 14-Pin Plastic DIP and Hermetic Cerdip Packages and in Chip Form MIL-STD-883B Processing Available Standard Military Drawing Available APPLICATIONS Active Filters Quad Output Buffers for 12- and 14-Bit DACs Input Buffers for Precision ADCs Photo Diode Preamplifier Applications PRODUCT DESCRIPTION
Quad Precision, Low Cost, High Speed, BiFET Op Amp AD713
CONNECTION DIAGRAMS Plastic (N) and Cerdip (Q) Packages
OUTPUT OUTPUT -IN +IN +VS +IN -IN OUTPUT 1 2 3 4 5 6 7 2 3 1 4 14 OUTPUT 13 -IN 12 +IN -IN +IN +VS +IN -IN OUTPUT NC
SOIC (R) Package
1 2 3 4 5 6 7 8 2 3 1 4 16 15 14 OUTPUT -IN +IN -VS +IN -IN OUTPUT NC
AD713
(TOP VIEW)
13 12 11 10 9
AD713
(TOP VIEW)
11 -V S 10 +IN 9 8 -IN
OUTPUT
NC = NO CONNECT
The AD713 is offered in a 16-pin SOIC, 14-pin plastic DIP and hermetic cerdip package, or in chip form.
PRODUCT HIGHLIGHTS
The AD713 is a quad operational amplifier, consisting of four AD711 BiFET op amps. These precision monolithic op amps offer excellent dc characteristics plus rapid settling times, high slew rates, and ample bandwidths. In addition, the AD713 provides the close matching ac and dc characteristics inherent to amplifiers sharing the same monolithic die. The single-pole response of the AD713 provides fast settling: l s to 0.01%. This feature, combined with its high dc precision, makes it suitable for use as a buffer amplifier for 12- or 14-bit DACs and ADCs. It is also an excellent choice for use in active filters in 12-, 14- and 16-bit data acquisition systems. Furthermore, the AD713's low total harmonic distortion (THD) level of 0.0003% and very close matching ac characteristics make it an ideal amplifier for many demanding audio applications. The AD713 is internally compensated for stable operation at unity gain and is available in seven performance grades. The AD713J and AD713K are rated over the commercial temperature range of 0C to +70C. The AD713A and AD713B are rated over the industrial temperature of -40C to +85C. The AD713S and AD713T are rated over the military temperature range of -55C to +125C and are available processed to MIL-STD-883B, Rev. C.
1. The AD713 is a high speed BiFET op amp that offers excellent performance at competitive prices. It upgrades the performance of circuits using op amps such as the TL074/ TL084, LT1058, LF347 and OPA404. 2. Slew rate is 100% tested for a guaranteed minimum of 16 V/s (J, A and S Grades). 3. The combination of Analog Devices' advanced processing technology, laser wafer drift trimming and well-matched ion-implanted JFETs provides outstanding dc precision. Input offset voltage, input bias current and input offset current are specified in the warmed-up condition and are 100% tested. 4. Very close matching of ac characteristics between the four amplifiers makes the AD713 ideal for high quality active filter applications.
REV. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703
AD713-SPECIFICATIONS (V =
S
15 V @ TA = +25 C unless otherwise noted)
AD713J/A/S Typ 0.3 0.5 5 95 95 15 40 55 10 Max 1.5 2/2/2 84 84 150 3.4/9.6/154 200 75 1.7/4.8/77 1.8 2.3/2.3/2.3 100 -130 -95 3.4 18 1.2 Min AD713K/B/T Typ 0.2 0.4 5 100 100 15 40 55 10 Max 0.5 0.7/0.7/1.0 20/20/15 Units mV mV V/C dB dB V/Month pA nA pA pA nA mV mV V/C pA dB dB MHz kHz V/s s %
Parameter INPUT OFFSET VOLTAGE Initial Offset Offset vs. Temp vs. Supply vs. Supply Long-Term Stability INPUT BIAS CURRENT2
1
Conditions
Min
TMIN to TMAX 78 76/76/76
TMIN to TMAX VCM = 0 V VCM = 0 V @ TMAX VCM = 10 V VCM = 0 V VCM = 0 V @ TMAX
75 1.7/4.8/77 120 35 0.8/2.2/36 0.8 1.0/1.0/1.3 25 35 -130 -95
INPUT OFFSET CURRENT
MATCHING CHARACTERISTICS Input Offset Voltage Input Offset Voltage TMIN to TMAX Input Offset Voltage Drift Input Bias Current Crosstalk f = 1 kHz f = 100 kHz FREQUENCY RESPONSE Small Signal Bandwidth Full Power Response Slew Rate Settling Time to 0.01% Total Harmonic Distortion INPUT IMPEDANCE Differential Common Mode INPUT VOLTAGE RANGE Differential3 Common-Mode Voltage4 Common Mode Rejection Ratio TMIN to TMAX VCM = 10 V TMIN to TMAX VCM = 11 V TMIN to TMAX 0.1 Hz to 10 Hz f = 10 Hz f = 100 Hz f = 1 kHz f = 10 kHz f = 1 kHz VO = 10 V; RL 2 k TMIN to TMAX RL 2 k TMIN to TMAX Short Circuit -11 78 76/76/76 72 70/70/70 Unity Gain VO = 20 V p-p Unity Gain f = 1 kHz; RL 2 k; VO = 3 V rms 3.0 16
0.5 0.7 8 10
0.4 0.6 6 10
4.0 200 20 1.0 0.0003
4.0 200 20 1.0 0.0003
1.2
3x1012 5.5 3x1012 5.5 20 +14.5, -11.5 +13 88 84 84 80 2 45 22 18 16 0.01 150 400 100/100/100 +13, -12.5 +13.9, -13.3 12/ 12/ 12 +13.8, -13.1 25 15 4.5 10.0 # of Transistors 120 18 13.5 4.5 200 100 -11 84 82 78 74
3x1012 5.5 3x1012 5.5 20 +14.5, -11.5 +13 94 90 90 84 2 45 22 18 16 0.01 400
pF pF V V V dB dB dB dB V p-p nV/Hz nV/Hz nV/Hz nV/Hz pA/Hz V/mV V/mV V V mA V V mA
INPUT VOLTAGE NOISE
INPUT CURRENT NOISE OPEN-LOOP GAIN OUTPUT CHARACTERISTICS Voltage Current POWER SUPPLY Rated Performance Operating Range Quiescent Current TRANSISTOR COUNT
+13, -12.5 +13.9, -13.3 12 +13.8, -13.1 25 15 10.0 120 18 12.0
NOTES 1 Input Offset Voltage specifications are guaranteed after 5 minutes of operation at T A = +25C. 2 Bias Current specifications are guaranteed maximum at either input after 5 minutes of operation at T A = +25C. For higher temperatures, the current doubles every 10C. 3 Defined as voltage between inputs, such that neither exceeds 10 V from ground. 4 Typically exceeding -14.1 V negative common-mode voltage on either input results in an output phase reversal. Specifications subject to change without notice.
-2-
REV. B
AD713
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Internal Power Dissipation2 Input Voltage3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V Output Short Circuit Duration (For One Amplifier) . . . . . . . . . . . . . . . . . . . . . . . . Indefinite Differential Input Voltage . . . . . . . . . . . . . . . . . . +VS and -VS Storage Temperature Range (Q) . . . . . . . . . . -65C to +150C Storage Temperature Range (N, R) . . . . . . . . -65C to +125C Operating Temperature Range AD713J/K . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0C to +70C AD713A/B . . . . . . . . . . . . . . . . . . . . . . . . . . -40C to +85C AD713S/T . . . . . . . . . . . . . . . . . . . . . . . . . -55C to +125C Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300C
NOTES 1 Stresses above those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Thermal Characteristics: 14-Pin Plastic Package: JC = 30C/Watt; JA = 100C/Watt 14-Pin Cerdip Package: JC = 30C/Watt; JA = 110C/Watt 16-Pin SOIC Package: JC = 30C/Watt; JA = 100C/Watt 3 For supply voltages less than 18 V, the absolute maximum input voltage is equal to the supply voltage.
ABSOLUTE MAXIMUM RATINGS 1, 2
ORDERING GUIDE
Model AD713AQ AD713BQ AD713JCHIPS AD713JN AD713JR-16 AD713JR-16-REEL AD713JR-16-REEL7 AD713KN AD713SCHIPS AD713SQ AD713SQ/883B AD713TQ AD713TQ/883B 5962-9063301MCA 5962-9063302MCA Temperature Range -40C to +85C -40C to +85C 0C to +70C 0C to +70C 0C to +70C 0C to +70C 0C to +70C 0C to +70C -55C to +125C -55C to +125C -55C to +125C -55C to +125C -55C to +125C -55C to +125C -55C to +125C Package Description 14-Pin Ceramic DIP 14-Pin Ceramic DIP Bare Die 14-Pin Plastic DIP 16-Pin Plastic SOIC 16-Pin Plastic SOIC 16-Pin Plastic SOIC 14-Pin Plastic DIP Bare Die 14-Pin Ceramic DIP 14-Pin Ceramic DIP 14-Pin Ceramic DIP 14-Pin Ceramic DIP 14-Pin Ceramic DIP 14-Pin Ceramic DIP Package Option* Q-14 Q-14 N-14 R-16 R-16 R-16 N-14 Q-14 Q-14 Q-14 Q-14 Q-14 Q-14
*N = Plastic DIP; Q = Cerdip; R = Small Outline IC (SOIC).
METALIZATION PHOTOGRAPH
Dimensions shown in inches and (mm). Contact factory for latest dimensions.
REV. B
-3-
AD713-Typical Characteristics
Figure 1. Input Voltage Swing vs. Supply Voltage
Figure 2. Output Voltage Swing vs. Supply Voltage
Figure 3. Output Voltage Swing vs. Load Resistance
Figure 4. Quiescent Current vs. Supply Voltage
Figure 5. Input Bias Current vs. Temperature
Figure 6. Output Impedance vs. Frequency, G = 1
Figure 7. Input Bias Current vs. Common Mode Voltage
Figure 8. Short Circuit Current Limit vs. Temperature
Figure 9. Gain Bandwidth Product vs. Temperature
-4-
REV. B
AD713
Figure 10. Open-Loop Gain and Phase Margin vs. Frequency
Figure 11. Open-Loop Gain vs. Supply Voltage
Figure 12. Power Supply Rejection vs. Frequency
Figure 13. Common Mode Rejection vs. Frequency
Figure 14. Large Signal Frequency Response
Figure 15. Output Swing and Error vs. Settling Time
Figure 16. Total Harmonic Distortion vs. Frequency
Figure 17. Input Noise Voltage Spectral Density
Figure 18. Slew Rate vs. Input Error Signal
REV. B
-5-
AD713
Figure 19. Crosstalk Test Circuit
Figure 20. Crosstalk vs. Frequency
Figure 21a. Unity Gain Follower
Figure 21b. Unity Gain Follower Pulse Response (Large Signal)
Figure 21c. Unity Gain Follower Pulse Response (Small Signal)
Figure 22a. Unity Gain Inverter
Figure 22b. Unity Gain Inverter Pulse Response (Large Signal)
Figure 22c. Unity Gain Inverter Pulse Response (Small Signal)
-6-
REV. B
AD713
MEASURING AD713 SETTLING TIME
The photos of Figures 24 and 25 show the dynamic response of the AD713 while operating in the settling time test circuit of Figure 23. The input of the settling time fixture is driven by a flat-top pulse generator. The error signal output from the false summing node of A1, the AD713 under test, is clamped, amplified by op amp A2 and then clamped again.
The error signal is thus clamped twice: once to prevent overloading amplifier A2 and then a second time to avoid overloading the oscilloscope preamp. A Tektronix oscilloscope preamp type 7A26 was carefully chosen because it recovers from the approximately 0.4 volt overload quickly enough to allow accurate measurement of the AD713's 1 s settling time. Amplifier A2 is a very high speed FET input op amp; it provides a voltage gain of 10, amplifying the error signal output of the AD713 under test (providing an overall gain of 5).
Figure 25. Settling Characteristics to -10 V Step. Upper Trace: Output of AD713 Under Test (5 V/div). Lower Trace: Amplified Error Voltage (0.01%/ div)
POWER SUPPLY BYPASSING
Figure 23. Settling Time Test Circuit
The power supply connections to the AD713 must maintain a low impedance to ground over a bandwidth of 4 MHz or more. This is especially important when driving a significant resistive or capacitive load, since all current delivered to the load comes from the power supplies. Multiple high quality bypass capacitors are recommended for each power supply line in any critical application. A 0.1 F ceramic and a 1 F electrolytic capacitor as shown in Figure 26 placed as close as possible to the amplifier (with short lead lengths to power supply common) will assure adequate high frequency bypassing in most applications. A minimum bypass capacitance of 0.1 F should be used for any application.
Figure 24. Settling Characteristics 0 V to +10 V Step. Upper Trace: Output of AD713 Under Test (5 V/div). Lower Trace: Amplified Error Voltage (0.01%/div)
Figure 26. Recommended Power Supply Bypassing
REV. B
-7-
AD713
A HIGH SPEED INSTRUMENTATION AMPLIFIER CIRCUIT A HIGH SPEED FOUR OP AMP CASCADED AMPLIFIER CIRCUIT
The instrumentation amplifier circuit shown in Figure 27 can provide a range of gains from unity up to 1000 and higher using only a single AD713. The circuit bandwidth is 1.2 MHz at a gain of 1 and 250 kHz at a gain of 10; settling time for the entire circuit is less than 5 s to within 0.01% for a 10 volt step, (G = 10). Other uses for amplifier A4 include an active data guard and an active sense input.
Figure 29 shows how the four amplifiers of the AD713 may be connected in cascade to form a high gain, high bandwidth amplifier. This gain of 100 amplifier has a -3 dB bandwidth greater than 600 kHz.
Figure 29. A High Speed Four Op Amp Cascaded Amplifier Circuit
Figure 27. A High Speed Instrumentation Amplifier Circuit
Table I provides a performance summary for this circuit. The photo of Figure 28 shows the pulse response of this circuit for a gain of 10.
Table I. Performance Summary for the High Speed Instrumentation Amplifier Circuit Gain RG Bandwidth T Settle (0.01%)
1 2 10
NC 20 k 4.04 k
1.2 MHz 1.0 MHz 0.25 MHz
2 s 2 s 5 s
Figure 30. THD Test Circuit
HIGH SPEED OP AMP APPLICATIONS AND TECHNIQUES DAC Buffers (I-to-V Converters)
Figure 28. The Pulse Response of the High Speed Instrumentation Amplifier. Gain = 10
The wide input dynamic range of JFET amplifiers makes them ideal for use in both waveform reconstruction and digital-audio DAC applications. The AD713, in conjunction with the AD1860 DAC, can achieve 0.0016% THD (here at a 4fs or a 176.4 kHz update rate) without requiring the use of a deglitcher. Just such a circuit is shown in Figure 31. The 470 pF feedback capacitor used with IC2a, along with op amp IC2b and its associated components, together form a 3-pole low-pass filter. Each or all of these poles can be tailored for the desired attenuation and phase characteristics required for a particular application. In this application, one half of an AD713 serves each channel in a twochannel stereo system. -8- REV. B
AD713
Figure 31. A D/A Converter Circuit for Digital Audio
Figure 33. The AD713 as an ADC Buffer Figure 32. Harmonic Distortion as Frequency for the Digital Audio Circuit of Figure 31
Driving the Analog Input of an A/D Converter
An op amp driving the analog input of an A/D converter, such as that shown in Figure 33, must be capable of maintaining a constant output voltage under dynamically changing load conditions. In successive approximation converters, the input current is compared to a series of switched trial currents. The comparison point is diode clamped but may vary by several hundred millivolts, resulting in high frequency modulation of the A/D input current. The output impedance of a feedback amplifier is made artificially low by its loop gain. At high frequencies, where the loop gain is low, the amplifier output impedance can approach its open loop value.
Most IC amplifiers exhibit a minimum open loop output impedance of 25 , due to current limiting resistors. A few hundred microamps reflected from the change in converter loading can introduce errors in instantaneous input voltage. If the A/D conversion speed is not excessive and the bandwidth of the amplifier is sufficient, the amplifier's output will return to the nominal value before the converter makes its comparison. However, many amplifiers have relatively narrow bandwidths, yielding slow recovery from output transients. The AD713 is ideally suited as a driver for A/D converters since it offers both a wide bandwidth and a high open loop gain.
REV. B
-9-
AD713
Figure 34. Buffer Recovery Time Source Current = 2 mA
Figure 37. Transient Response, RL = 2 k, CL = 500 pF
CMOS DAC APPLICATIONS
The AD713 is an excellent output amplifier for CMOS DACs. It can be used to perform both 2 and 4 quadrant operation. The output impedance of a DAC using an inverted R-2R ladder approaches R for codes containing many "1"s, 3R for codes containing a single "1" and infinity for codes containing all zeros. For example, the output resistance of the AD7545 will modulate between 11 k and 33 k. Therefore, with the DAC's internal feedback resistance of 11 k, the noise gain will vary from 2 to 4/3. This changing noise gain modulates the effect of the input offset voltage of the amplifier, resulting in nonlinear DAC amplifier performance. The AD713, with its guaranteed 1.5 mV input offset voltage, minimizes this effect achieving 12-bit performance. Figures 38 and 39 show the AD713 and a 12-bit CMOS DAC, the AD7545, configured for either a unipolar binary (2-quadrant multiplication) or bipolar (4-quadrant multiplication) operation. Capacitor C1 provides phase compensation which reduces overshoot and ringing.
Figure 35. Buffer Recovery Time Sink Current = 1 mA
The circuit of Figure 36 employs a 100 isolation resistor which enables the amplifier to drive capacitive loads exceeding 1500 pF; the resistor effectively isolates the high frequency feedback from the load and stabilizes the circuit. Low frequency feedback is returned to the amplifier summing junction via the low pass filter formed by the 100 series resistor and the load capacitance, C1. Figure 37 shows a typical transient response for this connection.
Driving A Large Capacitive Load
Figure 38. Unipolar Binary Operation
Figure 36. Circuit for Driving a Large Capacitance Load
Table II. Recommended Trim Resistor Values vs. Grades for AD7545 for VD = +5 V Trim Resistor JN/AQ/ SD KN/BQ/ TD LN/CQ/ UD GLN/GCQ/ GUD
R1 R2
500 150
200 68
100 33
20 6.8 -10-
Figure 39. Bipolar Operation
REV. B
AD713
Figure 40. A Programmable State Variable Filter Circuit
FILTER APPLICATIONS A Programmable State Variable Filter
For the state variable or universal filter configuration of Figure 40 to function properly, DACs A1 and B1 need to control the gain and Q of the filter characteristic, while DACs A2 and B2 must accurately track for the simple expression of fC to be true. This is readily accomplished using two AD7528 DACs and one AD713 quad op amp. Capacitor C3 compensates for the effects of op amp gain-bandwidth limitations. This filter provides low pass, high pass and band pass outputs and is ideally suited for applications where microprocessor control of filter parameters is required. The programmable range for component values shown is fC = 0 to 15 kHz and Q = 0.3 to 4.5.
GIC and FDNR FILTER APPLICATIONS
41 and 43 show the AD713 used in two typical active filters. The first shows a single AD713 simulating two coupled inductors configured as a one-third octave bandpass filter. A single section of this filter meets ANSI class II specifications and handles a 7.07 V rms signal with <0.002% THD (20 Hz-20 kHz). Figure 43 shows a 7-pole antialiasing filter for a 2 oversampling (88.2 kHz) digital audio application. This filter has <0.05 dB pass band ripple and 19.8 0.3 s delay, dc-20 kHz and will handle a 5 V rms signal (VS = 15 V) with no overload at any internal nodes. The filter of Figure 41 can be scaled for any center frequency by using the formula:
fC = 1.11 2RC
The closely matched and uniform ac characteristics of the AD713 make it ideal for use in GIC (gyrator) and FDNR (frequency dependent negative resistor) filter applications. Figures
where all resistors and capacitors scale equally. Resistors R3-R8 should not be greater than 2 k in value, to prevent parasitic oscillations caused by the amplifier's input capacitance.
Figure 41. A 1/3 Octave Filter Circuit
REV. B
-11-
AD713
If this is not practical, small lead capacitances (10-20 pF) should be added across R5 and R6. Figures 42 and 44 show the output amplitude vs. frequency of these filters.
Figure 42. Output Amplitude vs. Frequency of 1/3 Octave Filter
Figure 44. Relative Output Amplitude vs. Frequency of Antialiasing Filter
Figure 43. An Antialiasing Filter
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
14-Pin Plastic (N-14A) DIP Package
14-Pin Cerdip (Q-14) Package
16-Pin SOIC (R-16) Package
0.2992 (7.60) 0.2914 (7.40) 0.4193 (10.65) 0.3937 (10.00) 1 8
PIN 1
0.4133 (10.50) 0.3977 (10.00)
0.1043 (2.65) 0.0926 (2.35)
0.0118 (0.30) 0.0040 (0.10)
0.0500 (1.27) BSC
0.0192 (0.49) 0.0138 (0.35)
0.0291 (0.74) x 45 0.0098 (0.25)
0.0125 (0.32) 0 0091 (0 23)
8 0
0.0500 (1.27) 0.0157 (0.40)
-12-
REV. B
PRINTED IN U.S.A.
16
9
C1206a-5-11/90

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